Switching power supply for 2161 840. Switching power supply for umzch on ir2161. Features of laboratory blocks

Do-it-yourself homemade switching power supply.

The author of the design (Sergey Kuznetsov, his website is classd.fromru.com) developed this homemade network power supply
for powering a powerful UMZCH (Audio Frequency Power Amplifier). Advantages of switching mains power supplies before conventional transformer power supplies are obvious:

• The weight of the resulting product is much lower
• The dimensions of the switching power supply are much smaller.
• The efficiency of the product and, accordingly, the heat dissipation is lower
• The range of supply voltages (voltage surges in the network) at which the power supply can operate stably is much wider.

However, manufacturing a switching mains power supply requires much more effort and knowledge compared to manufacturing a conventional low-frequency 50 Hz power supply. A low-frequency power supply consists of a mains transformer, a diode bridge and smoothing filter capacitors, while a pulsed one has a much more complex structure.

The main disadvantage of switching network power supplies is the presence of high-frequency interference, which will have to be overcome if the printed circuit board is routed incorrectly, or if the component base is incorrectly selected. When you turn on the UPS, as a rule, there is a strong spark at the outlet. This is due to the large peak startup current of the power supply, due to the charge of the input filter capacitors. To eliminate such current surges, developers are designing various “soft start” systems that, in the first phase of operation, charge the filter capacitors with a low current, and when the charging is complete, they organize the supply of full mains voltage to the UPS. In this case, a simplified version of such a system is used, which is a series-connected resistor and thermistor that limits the charging current of the capacitors.

The circuit is based on the IR2153 PWM controller in a standard connection circuit. Field-effect transistors IRFI840GLC can be replaced with IRFIBC30G; the author does not recommend installing other transistors, since this will entail the need to reduce the ratings of R2, R3 and, accordingly, an increase in the heat generated. The voltage on the PWM controller must be at least 10 Volts. It is desirable for the microcircuit to operate at a voltage of 11-14 Volts. Components L1 C13 R8 improve the operating mode of transistors.

The chokes located at the output of the 10 μg power source are wound with 1 mm wire on ferrite dumbbells with a magnetic permeability of 600 NN. You can wind it on rods from old receivers, 10-15 turns are enough. Capacitors in the power supply must be low-impedance in order to reduce RF noise.

The transformer was calculated using the Transformer 2 program. The induction should be chosen as low as possible, preferably no more than 0.25. The frequency is around 40-80k. The author does not recommend the use of domestically produced rings, due to the non-identity of ferrite parameters and significant losses in the transformer. The printed circuit board was designed for a transformer of standard size 30x19x20. When setting up the power supply, it is prohibited to connect the ground of the oscilloscope to the connection point of the transistors. It is advisable to start the power supply for the first time with a 220V lamp with a power of 25-40W connected in series with the source, and the UPS should not be heavily loaded. The printed circuit board of the block in LAY format can be downloaded or

This article is devoted to the 2161 Second Edition (SE) series of switching power supplies based on the IR2161 controller.

Here we will talk about three completed SMPS based on IR2161, each of which will be better than the previous one, their circuits, printed circuit boards will be given, and some important points will be described.

But before we begin the story directly about the power supplies themselves, I would like to dwell on the IR2161 itself and describe in detail the principle and features of its operation. As time has shown, even those who assemble their own 2161 pulse blocks have little idea how this chip works (China+, hello). It is for this reason that you can come across a lot of elementary questions to which you could easily find answers in the datasheet, but apparently not everyone is able to understand the material presented there, and many are simply too lazy to delve into it.

IR2161 is a specialized intelligent half-bridge converter integrated circuit for halogen lamps (electronic transformer). Remember Toshibra electronic transformers? It is for such “electronic transformers” that this controller was developed, but not for those cheap Chinese fakes like Toshibra, but for good and high-quality electronic transformers that have nothing in common with the depicted Toshibra.

The IR2161 controller includes all the necessary protections, and also allows the converter to be adapted for dimming with a standard phase control dimer (the ability to dim for our purposes does not matter). There is also compensation of the output voltage depending on the power consumed by the load. The IR2161 features adaptive dead time that improves operating stability and frequency modulation "dither" to reduce electromagnetic radiation (EMR). All this is integrated into a small 8-pin DIP or SOIC package, allowing the size of the SMPS to be kept as small as possible.

Let me briefly list IR2161 capabilities listed in the datasheet:

• Short circuit and overload protection;
• Auto reset short circuit protection;
• Frequency modulation "dither" (to reduce EMI);
• Microcurrent startup (for initial startup of the controller, a current of no more than 300 μA is sufficient);
• Possibility of dimming (but we are not interested in this);
• Output voltage compensation (a kind of voltage stabilization);
• Soft start;
• Adaptive dead time ADT;
• Compact body;
• Produced using lead-free technology (Leed-Free).

I will give some important ones for us technical specifications:

Maximum inflow/outflow current: ±500mA
A sufficiently large current allows you to control powerful switches and build quite powerful switching power supplies based on this controller without the use of additional drivers;

Maximum current consumed by the controller: 10mA
Based on this value, the power circuits of the microcircuit are designed;

Minimum operating voltage of the controller: 10.5V
At a lower supply voltage, the controller switches to UVLO mode and the oscillation stops;

Minimum stabilization voltage of the zener diode built into the controller: 14.5V
The external zener diode must have a stabilization voltage no higher than this value to avoid damage to the microcircuit due to shunting excess current to the COM pin;

Voltage at the CS pin to trigger overload protection: 0.5V
The minimum voltage at the CS pin at which the overload protection is triggered;

Voltage at the CS pin for short circuit protection: 1V
The minimum voltage at the CS pin at which short circuit protection is triggered;

Operating frequency range: 34 - 70 kHz
The operating frequency is not directly set and depends only on the power consumed by the load;

Default dead time: 1µS
Used when it is impossible to work in adaptive dead time (ADT) mode, as well as when there is no load;

Operating frequency in soft start mode: 130 kHz
The frequency at which the controller operates in soft start mode;

The main attention should now be paid to what operating modes of the microcircuit exist and in what sequence they are located relative to each other. I will focus on describing the operating principle of each of the circuit blocks, and I will describe the sequence of their operation and the conditions for transition from one mode to another more briefly. I'll start with a description of each of the blocks of the diagram:

Under-voltage Lock-Out Mode (UVLO)- the mode in which the controller is when its supply voltage is below the minimum threshold value (approximately 10.5V).

Soft Start Mode- operating mode in which the controller oscillator operates at an increased frequency for a short time. When the oscillator is turned on, its operating frequency is initially very high (about 130 kHz). This causes the converter output voltage to be lower because the power supply transformer has a fixed inductance which will have a higher impedance at higher frequency and thus reduces the voltage on the primary winding. Reduced voltage will naturally result in reduced current in the load. As the CSD capacitor charges from 0 to 5V, the oscillation frequency will gradually decrease from 130 kHz to the operating frequency. The duration of the soft start sweep will depend on the capacitance of the CSD capacitor. However, since the CSD capacitor also sets the shutdown delay time and participates in the operation of the voltage compensation unit, its capacitance must be strictly 100nF.

Soft start problem. I would like to be completely honest and mention the fact that if there are high-capacity filter capacitors at the output of the power supply, soft start most often does not work and the SMPS starts immediately at the operating frequency, bypassing the soft start mode. This happens due to the fact that at the moment of start, the discharged capacitors in the secondary circuit have a very low intrinsic resistance and a very high current is required to charge them. This current causes short-circuit protection to operate briefly, after which the controller immediately restarts and goes into RUN mode, bypassing the soft start mode. You can combat this by increasing the inductance of the chokes in the secondary circuit, located immediately after the rectifier. Chokes with high inductance extend the charging process of the output filter capacitors; in other words, the capacitors are charged with a smaller current, but longer in time. A lower charging current does not trigger the protection at start and allows the soft start to perform its functions normally. Just in case, regarding this issue, I contacted the manufacturer’s technical support, to which I received the following answer:

"A typical halogen converter has an AC output without rectifiers or output capacitors. Soft starting works by reducing the frequency. To achieve soft starting, the transformer needs to have significant leakage. However, this should be possible in your case. Try placing the inductor on the secondary side of the bridges diodes to the capacitor.

Best wishes.
Infineon Technologies
Steve Rhyme, Support Engineer"

My assumptions about the reason for the unstable operation of soft start turned out to be correct and, moreover, they even offered me the same method of dealing with this problem. And again, to be completely honest, it should be added that the use of coils with increased inductance, relative to those usually used at the output of the SMPS, improves the situation, but does not completely eliminate the problem. However, this problem can be tolerated given that there is a thermistor at the SMPS input that limits the inrush current.

Run Mode, operating mode. When the soft start is completed, the system enters voltage compensated operating mode. This function provides some stabilization of the converter output voltage. Voltage compensation occurs by changing the operating frequency of the converter (increasing the frequency reduces the output voltage), although the accuracy of this type of “stabilization” is not high, it is nonlinear and depends on many parameters and, therefore, is not easy to predict. IR2161 monitors the load current through a current resistor (RCS). The peak current is detected and amplified in the controller and then applied to the CSD pin. The voltage on the CSD capacitor, in operating mode (voltage compensation mode), will vary from 0 (at minimum load) to 5V (at maximum load). In this case, the generator frequency will vary from 34 kHz (Vcsd = 5V) to 70 kHz (Vcsd = 0V).

It is also possible to attach feedback to the IR2161, which will allow you to organize almost complete stabilization of the output voltage and will allow you to much more accurately monitor and maintain the required voltage at the output:

We will not consider this scheme in detail within the framework of this article.

Shut Down Mode, shutdown mode. The IR2161 contains a two-position automatic shutdown system that detects both short circuit and overload conditions of the inverter. The voltage at the CS pin is used to determine these conditions. If the output of the converter is shorted, a very large current will flow through the switches and the system must shut down within a few periods of time on the grid, otherwise the transistors will be quickly destroyed due to thermal runaway of the junction. The CS pin has a turn-off delay to prevent nuisance tripping, either due to inrush current at turn-on or due to transient currents. Lower threshold (when Vcs > 0.5< 1 В), имеет намного большую задержку до отключения ИИП. Задержка для отключения по перегрузке приблизительно равна 0,5 сек. Оба режима отключения (по перегрузке и по короткому замыканию), имеют автоматический сброс, что позволяет контроллеру возобновить работу примерно через 1 сек после устранения перегрузки или короткого замыкания. Это значит, что если неисправность будет устранена, преобразователь может продолжить нормально работать. Осциллятор работает на минимальной рабочей частоте (34 кГц), когда конденсатор CSD переключается к цепи отключения. В режиме плавного пуска или рабочем режиме, если превышен порог перегрузки (Vcs >0.5V), IR2161 quickly charges CSD up to 5V. When the voltage at the CS pin is greater than 0.5V and when the short circuit threshold of 1V is exceeded, the CSD will charge from 5V to the controller supply voltage (10-15V) in 50ms. When the overload threshold voltage Vcs is more than 0.5V but less than 1V, the CSD is charged from 5V to the supply voltage in approximately 0.5 sec. It should be remembered and taken into account the fact that high-frequency pulses with a 50% duty cycle and a sinusoidal envelope appear at the CS pin - this means that only at the peak of the network voltage the CSD capacitor will be charged in stages, in each half-cycle. When the voltage on the CSD capacitor reaches the supply voltage, the CSD is discharged to 2.4V and the converter starts again. If the fault is still present, the CSD starts charging again. If the fault disappears, the CSD will discharge to 2.4V, and then the system will automatically return to the voltage compensation operating mode.

STANDBY mode, standby mode- the mode in which the controller is in the case of insufficient supply voltage, while it consumes no more than 300 μA. In this case, the oscillator is naturally turned off and the SMPS does not work; there is no voltage at its output.

Blocks Fault Timing Mode, Delay and Fault Mode, although shown in the block diagram, are not essentially operating modes of the controller; rather, they can be attributed to transition stages (Delay and Fault Mode) or conditions for transition from one mode to another (Fault Timing Mode).

Now I’ll describe how does it all work together:
When power is applied, the controller starts in UVLO mode. As soon as the controller supply voltage exceeds the minimum voltage value required for stable operation, the controller switches to soft start mode, the oscillator starts at a frequency of 130 kHz. The CSD capacitor charges smoothly up to 5V. As the external capacitors charge, the operating frequency of the oscillator decreases to the operating frequency. Thus, the controller switches to RUN mode. As soon as the controller enters RUN mode, the CSD capacitor is instantly discharged to ground potential and is connected by an internal switch to the voltage compensation circuit. If the SMPS starts not at idle, but under load, there will be a potential at the CS pin proportional to the load value, which, through the internal circuits of the controller, will affect the voltage compensation unit and will prevent the CSD capacitor from being completely discharged after the soft start is completed. Thanks to this, the start will not occur at the maximum frequency of the operating range, but at a frequency corresponding to the load value at the output of the SMPS. After switching to RUN mode, the controller works according to the situation: either it remains working in this mode until you get tired and unplug the power supply from the outlet, or... In case of overheating, the controller goes into FAULT mode, the oscillator stops working . After the chip cools down, a restart occurs. In the event of an overload or short circuit, the controller goes into Fault Timing mode, and the external capacitor CSD is instantly disconnected from the voltage compensation unit and connected to the shutdown unit (the CSD capacitor in this case sets the controller shutdown delay time). The operating frequency is instantly reduced to the minimum. In case of overload (when the voltage at the CS pin > 0.5< 1 В), контроллер переходит в режим SHUTDOWN и выключается, но происходит это не мгновенно, а только в том случае, если перегрузка продолжается дольше половины секунды. Если перегрузки носят импульсный характер с продолжительностью импульса не более 0,5 сек, то контроллер будет просто работать на минимально возможно частоте, постоянно переключаясь между режимами RUN, Fault Timing, Delay, RUN (при этом будут отчетливо слышны щелчки). Когда напряжение на выводе CS превышает 1В, срабатывает защита от короткого замыкания. При устранении перегрузки или короткого замыкания, контроллер переходит в режим STANDBY и при наличии благоприятных условий для перезапуска, минуя режим софт-старта, переходит в режим RUN.

Now that you understand how the IR2161 works (I hope so), I will tell you about the switching power supplies themselves based on it. I want to immediately warn you that if you decide to assemble a switching power supply based on this controller, then you should assemble the SMPS guided by the latest, most advanced circuit on the corresponding printed circuit board. Therefore, the list of radio elements at the bottom of the article will be given only for the latest version of the power supply. All intermediate editions of the IIP are shown only to demonstrate the process of improving the device.

And the first IIP that will be discussed is conventionally named by me 2161 SE 2.

The main and key difference between the 2161 SE 2 and the SMPS described in the first article is the presence of a controller self-supply circuit, which made it possible to get rid of boiling quenching resistors and, accordingly, increase the efficiency by several percent. Other equally significant improvements were also made: optimization of the printed circuit board layout, more output terminals were added for connecting the load, and a varistor was added.

The SMPS diagram is shown in the image below:

The self-powering circuit is built on VD1, VD2, VD3 and C8. Due to the fact that the self-supply circuit is connected not to a low-frequency 220V network (with a frequency of 50Hz), but to the primary winding of a high-frequency transformer, the capacity of the self-supply quenching capacitor (C8) is only 330pF. If self-supply was organized from a low-frequency 50Hz network, then the capacity of the quenching capacitor would have to be increased 1000 times, and of course such a capacitor would take up much more space on the printed circuit board. The described method of self-powering is no less effective than self-powering from a separate winding of a transformer, but it is much simpler. Zener diode VD1 is necessary to facilitate the operation of the built-in zener diode of the controller, which is not capable of dissipating significant power and without installing an external zener diode can simply be broken, which will lead to a complete loss of functionality of the microcircuit. The stabilization voltage VD1 should be in the range of 12 - 14V and should not exceed the stabilization voltage of the controller's built-in zener diode, which is approximately 14.5V. As VD1, you can use a zener diode with a stabilization voltage of 13V (for example, 1N4743 or BZX55-C13), or use several zener diodes connected in series, which is what I did. I connected two zener diodes in series: one of them was 8.2V, the other was 5.1V, which ultimately gave a resulting voltage of 13.3V. With this approach to powering the IR2161, the controller’s supply voltage does not sags and is practically independent of the load size connected to the SMPS output. In this scheme, R1 is only needed to start the controller, so to speak, for the initial kick. R1 gets a little warm, but not nearly as much as it was in the first version of this power supply. The use of high-resistance resistor R1 provides another interesting feature: the voltage at the output of the SMPS does not appear immediately after being connected to the network, but after 1-2 seconds, when C3 is charged to the minimum voltage of 2161 (approximately 10.5V).

Starting with this SMPS and all subsequent ones, a varistor is used at the SMPS input; it is designed to protect the SMPS from exceeding the input voltage above the permissible value (in this case - 275V), and also very effectively suppresses high-voltage interference by preventing them from entering the SMPS input from network and without releasing interference from the SMPS back into the network.

In the rectifier of the secondary power supply of the power supply, I used SF54 diodes (200V, 5A) two in parallel. The diodes are located on two floors, the leads of the diodes should be as long as possible - this is necessary for better heat dissipation (the leads are a kind of radiator for the diode) and better air circulation around the diodes.

The transformer in my case is made on a core from a computer power supply - ER35/21/11. The primary winding has 46 turns in three 0.5mm wires, two secondary windings have 12 turns in three 0.5mm wires. The input and output chokes are also taken from the computer power supply.

The described power supply is capable of delivering 250W to the load for a long time (without operating time limitation), and 350W for a short time (no more than a minute). When using this SMPS in dynamic load mode (for example, to power an audio frequency power amplifier of class B or AB), it is possible to power an UMZCH with a total output power of 300W (2x150W in stereo mode) from this switching power supply.

Oscillogram on the primary winding of the transformer (without snubber, R5 = 0.15 Ohm, 190W output):

As can be seen from the oscillogram, with an output power of 190 W, the operating frequency of the SMPS is reduced to 38 kHz; at idle, the SMPS operates at a frequency of 78 kHz:

From the oscillograms, in addition, it is clearly visible that there are no outliers on the graph, and this undoubtedly characterizes this SMPS positively.

At the output of the power supply, in one of the arms you can see the following picture:

The ripple has a frequency of 100Hz and a ripple voltage of approximately 0.7V, which is comparable to the ripple at the output of a classic, linear, non-stabilized power supply. For comparison, here is an oscillogram taken when operating at the same output power for a classic power supply (capacitor capacity 15000 μF in the arm):

As can be seen from the oscillograms, the supply voltage ripple at the output of a switching power supply is lower than that of a classic power supply of the same power (0.7V for an SMPS, versus 1V for a classic unit). But unlike a classic power supply, a small high-frequency noise is noticeable at the output of the SMPS. However, there is no significant high-frequency interference or emissions. The ripple frequency of the supply voltage at the output is 100Hz and is caused by the voltage ripple in the primary circuit of the SMPS along the +310V bus. To further reduce ripple at the SMPS output, it is necessary to increase the capacitance of capacitor C9 in the primary circuit of the power supply or the capacitance of the capacitors in the secondary circuit of the power supply (the former is more effective), and to reduce high-frequency interference, use chokes with higher inductance at the SMPS output.

The PCB looks like this:

The following SMPS diagram that will be discussed is 2161 SE 3:

The finished power supply assembled according to this diagram looks like this:

There are no fundamental differences in the circuit from SE 2; the differences mainly concern the printed circuit board. The circuit added only snubbers in the secondary windings of the transformer - R7, C22 and R8, C23. The values ​​of the gate resistors have been increased from 22 Ohm to 51 Ohm. The value of capacitor C4 has been reduced from 220 µF to 47 µF. Resistor R1 is assembled from four 0.5W resistors, which made it possible to reduce the heating of this resistor and make the design slightly cheaper because In my area, four half-watt resistors are cheaper than one two-watt one. But the opportunity to install one two-watt resistor remains. In addition, the value of the self-feeding capacitor was increased to 470pF, there was no particular point in this, but it was done as an experiment, the flight was normal. MUR1560 diodes in a TO-220 package are used as rectifier diodes in the secondary circuit. Optimized and reduced printed circuit board. The dimensions of the SE 2 printed circuit board are 153x88, while the SE 3 printed circuit board has dimensions of 134x88. The PCB looks like this:

The transformer is made on a core from a computer power supply - ER35/21/11. The primary winding has 45 turns in three 0.5mm wires, two secondary windings have 12 turns in four 0.5mm wires. The input and output chokes are also taken from the computer power supply.

The very first inclusion of this SMPS in the network showed that the snubbers in the secondary circuit of the power supply were clearly superfluous; they were immediately soldered off and were not used further. Later the snubber of the primary winding was also soldered off, as it turned out it did much more harm than good.

It was possible to extract 300-350W of power from this power supply for a long time; for a short time (no more than a minute) this SMPS can supply up to 500W; after a minute of operation in this mode, the overall radiator heats up to 60 degrees.

Look at the oscillograms:

Everything is still beautiful, the rectangle is almost perfectly rectangular, there are no outliers. With snubbers, oddly enough, everything was not so beautiful.

The following diagram is the final and most advanced 2161 SE 4:

When assembled, the device according to this diagram looks like this:

Like last time, there were no major changes in the scheme. Perhaps the most noticeable difference is that the snubbers have disappeared, both in the primary circuit and in the secondary ones. Because, as my experiments have shown, due to the peculiarities of the IR2161 controller, snubbers only interfere with its operation and are simply contraindicated. Other changes were also made. The values ​​of the gate resistors (R3 and R4) have been reduced from 51 to 33 Ohms. In series with the self-feeding capacitor C7, a resistor R2 is added to protect against overcurrents when charging capacitors C3 and C4. Resistor R1 still consists of four half-watt resistors, and resistor R6 is now hidden under the board and consists of three SMD resistors of the 2512 format. Three resistors provide the required resistance, but it is not necessary to use exactly three resistors; depending on the required power, you can use one, two or three resistors are acceptable. Thermistor RT1 has been moved from the SMPS to the +310V target. The remaining measurements concern only the layout of the printed circuit board and it looks like this:

A safety gap has been added to the printed circuit board between the primary and secondary circuits, and a through cut has been made in the board at the narrowest point.

The transformer is exactly the same as in the previous power supply: it is made on a core from a computer power supply - ER35/21/11. The primary winding has 45 turns in three 0.5mm wires, two secondary windings have 12 turns in four 0.5mm wires. The input and output chokes are also taken from the computer power supply.

The output power of the power supply remained the same - 300-350W in long-term mode and 500W in short-term mode (no more than a minute). From this SMPS you can power a UMZCH with a total output power of up to 400W (2x200W in stereo mode).

Now let's look at the oscillograms on the primary winding of the transformer of this switching power supply:

Everything is still beautiful: the rectangle is rectangular, there are no outliers.

At the output of one of the arms of the power supply, at idle, you can observe the following picture:

As you can see, the output contains negligible high-frequency noise with a voltage of no more than 8 mV (0.008 V).

Under load, at the output, you can observe the already well-known ripples with a frequency of 100 Hz:

With an output power of 250W, the ripple voltage at the output of the SMPS is 1.2V, which, considering the lower capacitance of the capacitors in the secondary circuit (2000uF in the shoulder, versus 3200uF for SE2) and the high output power at which the measurements were made, looks very good. The high-frequency component at a given output power (250W) is also insignificant, has a more ordered character and does not exceed 0.2V, which is a good result.

Setting the protection threshold. The threshold at which the protection will operate is set by the RCS resistor (R5 - in SE 2, R6 - in SE 3 and SE 4).

This resistor can be either output or SMD format 2512. RCS can be composed of several resistors connected in parallel.
The RCS denomination is calculated using the formula: Rcs = 32 / Pnom. Where, Pnom is the output power of the SMPS, above which the overload protection will operate.
Example: let's say that we need the overload protection to be triggered when the output power exceeds 275W. We calculate the resistor value: Rcs=32/275=0.116 Ohm. You can use either one 0.1 Ohm resistor, or two 0.22 Ohm resistors connected in parallel (which will result in 0.11 Ohm), or three 0.33 Ohm resistors, also connected in parallel (which will result in 0.11 Ohm) .

Now it’s time to touch on the topic that interests people the most - calculation of a transformer for a switching power supply. Due to your numerous requests, I will finally tell you in detail how to do this.

First of all, we need a core with a frame, or just a core if it is a ring-shaped core (shape R).

Cores and frames can be of completely different configurations and can be used in any way. I used an ER35 frame core from a computer power supply. The most important thing is that the core does not have a gap; cores with a gap cannot be used.

Next, we need a program to calculate the transformer; the Lite-CalcIT program is best suited for these purposes:

By default, immediately after starting the program, you will see similar numbers.
Starting the calculation, the first thing we will do is select the shape and dimensions of the core in the upper right corner of the program window. In my case, the shape is ER, and the sizes are 35/21/11.

The dimensions of the core can be measured independently; how to do this can be easily understood from the following illustration:

Next, select the core material. It’s good if you know what material your core is made of, if not, then it’s okay, just choose the default option - N87 Epcos. In our conditions, the choice of material will not have a significant impact on the final result.

The next step is to select the converter circuit; ours is half-bridge:

In the next part of the program - “supply voltage”, select “variable” and indicate 230V in all three windows.

In the “converter characteristics” part, we indicate the bipolar output voltage we need (voltage of one arm) and the required output power of the SMPS, as well as the diameter of the wire with which you want to wind the secondary and primary windings. In addition, the type of rectifier used is selected - “bipolar with a midpoint”. There we also check the box “use the desired diameters” and under “stabilization of outputs” select “no”. Select the type of cooling: active with a fan or passive without it. You should end up with something like this:

The actual values ​​of the output voltages will be greater than what you indicate in the program when calculating. In this case, with a voltage of 2x45V specified in the program, the output of a real SMPS will be approximately 2x52V, so when calculating, I recommend specifying a voltage that is 3-5V less than required. Or indicate the required output voltage, but wind one turn less than indicated in the program calculation results. The output power should not exceed 350W (for 2161 SE 4). The diameter of the wire for winding, you can use any one you have, you need to measure and indicate its diameter. You should not wind the windings with a wire with a diameter of more than 0.8 mm; it is better to wind the windings using several (two, three or more) thin wires than one thick wire.

After all this, click on the “calculate” button and get the result, in my case it turned out like this:

We focus our attention on the points highlighted in red. The primary winding in my case will consist of 41 turns, wound in two wires with a diameter of 0.5 mm each. The secondary winding consists of two halves of 14 turns, wound in three wires with a diameter of 0.5 mm each.

After receiving all the necessary calculation data, we proceed directly to winding the transformer.
It seems to me that there is nothing complicated here. I'll tell you how I do it. First, the entire primary winding is wound. One of the ends of the wire(s) is stripped and soldered to the corresponding terminal of the transformer frame. After which the winding begins. The first layer is wound and then a thin layer of insulation is applied. After which the second layer is wound and a thin layer of insulation is applied again and thus the entire required number of turns of the primary winding is wound. It is best to wind the windings turn to turn, but you can also do it askew or just “anyhow”, this will not play a noticeable role. After the required number of turns have been wound, the end of the wire(s) is cut off, the end of the wire is stripped and soldered to another corresponding terminal of the transformer. After winding the primary winding, a thick layer of insulation is applied to it. It is best to use a special Mylar tape as insulation:

The same tape is used to insulate the windings of pulse transformers of computer power supplies. This tape conducts heat well and has high heat resistance. From available materials, it is recommended to use: FUM tape, masking tape, paper plaster or a baking sleeve cut into long strips. It is strictly forbidden to use PVC and fabric insulating tape, stationery tape, or fabric plaster to insulate windings.

After the primary winding is wound and insulated, we proceed to winding the secondary winding. Some people wind two halves of the winding at once and then separate them, but I wind the halves of the secondary winding one by one. The secondary winding is wound in the same way as the primary. First, we strip and solder one end of the wire(s) to the corresponding terminal of the transformer frame, wind the required number of turns, applying insulation after each layer. Having wound the required number of turns of one half of the secondary winding, we strip and solder the end of the wire to the corresponding terminal of the frame and apply a thin layer of insulation. We solder the beginning of the wire of the next half of the winding to the same terminal as the end of the previous half of the winding. We wind in the same direction, the same number of turns as the previous half of the winding, applying insulation after each layer. Having wound the required number of turns, solder the end of the wire to the corresponding terminal of the frame and apply a thin layer of insulation. There is no need to apply a thick layer of insulation after winding the secondary winding. At this point, the winding can be considered complete.

After winding is completed, it is necessary to insert the core into the frame and glue the core halves together. For gluing, I use one-second super glue. The glue layer should be minimal so as not to create a gap between the parts of the core. If you have a ring core (shape R), then naturally you won’t have to glue anything, but the winding process will be less convenient and will take more effort and nerves. In addition, the ring core is less convenient due to the fact that you will have to create and mold the transformer leads yourself, as well as think about attaching the finished transformer to the printed circuit board.

Upon completion of winding and assembly of the transformer, you should get something like this:

For convenience of narration, I will also add here the SMPS 2161 SE 4 diagram for a brief description talk about the element base and possible replacements.

Let's go in order - from entrance to exit. At the input, the mains voltage meets fuse F1; the fuse can have a rating from 3.15A to 5A. Varistor RV1 must be designed for 275V, such a varistor will be marked 07K431, but it is also possible to use variators 10K431 or 14K431. It is also possible to use a varistor with a higher threshold voltage, but the effectiveness of protection and noise suppression will be noticeably lower. Capacitors C1 and C2 can be either regular film capacitors (such as CL-21 or CBB-21) or noise-suppressing type (for example X2) for a voltage of 275V. We unsolder the dual inductor L1 from a computer power supply or other faulty equipment. The inductor can be made independently by winding 20-30 turns on a small ring core, with a wire with a diameter of 0.5 - 0.8 mm. The VDS1 diode bridge can be any for a current from 6 to 8A, for example, indicated in the diagram - KBU08 (8A) or RS607 (6A). Any slow or fast diode with a current from 0.1 to 1A and a reverse voltage of at least 400V is suitable as VD4. R1 can consist of either four half-watt resistors of 82 kOhm, or be one two-watt resistor with the same resistance. Zener diode VD1 must have a stabilization voltage in the range of 13 - 14V; it is allowed to use either one zener diode or a series connection of two zener diodes with a lower voltage. C3 and C5 can be either film or ceramic. C4 should have a capacitance of no more than 47 µF, voltage 16-25V. Diodes VD2, VD3, VD5 must be very fast, for example - HER108 or SF18. C6 can be either film or ceramic. Capacitor C7 must be designed for a voltage of at least 1000V. C9 can be either film or ceramic. The R6 rating must be calculated for the required output power, as described above. As R6, you can use either SMD resistors of the 2512 format or output one- or two-watt resistors; in any case, the resistor(s) are installed under the board. Capacitor C8 must be film (type CL-21 or CBB-21) and have an allowable operating voltage of at least 400V. C10 is an electrolytic capacitor with a voltage of at least 400V; the magnitude of low-frequency ripples at the output of the SMPS depends on its capacitance. RT1 is a thermistor, you can buy it, or you can unsolder it from a computer power supply, its resistance should be from 10 to 20 Ohms and the permissible current should be at least 3A. Both the IRF740 indicated in the diagram and other transistors with similar parameters, for example, IRF840, 2SK3568, STP10NK60, STP8NK80, 8N60, 10N60, can be used as transistors VT1 and VT2. Capacitors C11 and C13 must be film (type CL-21 or CBB-21) with a permissible voltage of at least 400V, their capacitance must not exceed the 0.47 μF indicated in the diagram. C12 and C14 are ceramic, high-voltage capacitors for a voltage of at least 1000V. The VDS2 diode bridge consists of four diodes connected by a bridge. As VDS2 diodes, it is necessary to use very fast and powerful diodes, for example, such as - MUR1520 (15A, 200V), MUR1560 (15A, 600V), MUR820 (8A, 200V), MUR860 (8A, 600V), BYW29 (8A, 200V) , 8ETH06 (8A, 600V), 15ETH06 (15A, 600V). Chokes L2 and L3 are soldered from the computer power supply or made independently. They can be wound either on individual ferrite rods or on a common ring core. Each of the chokes should contain from 5 to 30 turns (more is better), with a wire with a diameter of 1 - 1.5 mm. Capacitors C15, C17, C18, C20 must be film (type CL-21 or CBB-21) with a permissible voltage of 63V or more, the capacitance can be any, the larger their capacitance, the better, the stronger the suppression of high-frequency interference. Each of the capacitors designated in the diagram as C16 and C19 consists of two 1000uF 50V electrolytic capacitors. In your case you may need to use higher voltage capacitors.

And as a final touch, I’ll show you a photo that shows the evolution of the switching power supplies I created. Each subsequent SMPS is smaller, more powerful and better quality than the previous one:

That's all! Thank you for your attention!

List of radioelements

Designation	Type	Denomination	Quantity	Note	Shop
	Switching Power Supply 2161 SE 4
R1	Resistor	82 kOhm	4	0.5W	Search in LCSC
R2	Resistor	4.7 Ohm	1	0.25W	Search in LCSC
R3, R4	Resistor	33 Ohm	2	0.25W	Search in LCSC
R5	Resistor	1 kOhm	1	0.25W	Search in LCSC
R6*	Resistor	0.47 Ohm	3	SMD 2512 or output 1-2W, rating calculated*	Search in LCSC
RT1	Resistor	10D-11	1	Thermistor, 10Ohm, 3A	Search in LCSC
RV1	Resistor	07K431	1	Varistor 275V	Search in LCSC
C1, C2	Capacitor	100 nF	2	X2 (275V) or CL-21 (400V)	Search in LCSC
C8	Capacitor	100 nF	1	CL-21 (400V) or СBB-21 (400V)	Search in LCSC
C3, C5	Capacitor	100 nF	2		Search in LCSC
C4	Capacitor	47 µF	1	Electrolytic 25V	Search in LCSC
C6	Capacitor	220 nF	1	CL-11 (100V) or K10-17 (50V)	Search in LCSC
C7	Capacitor	330 pF	1	CT-81 (1000V) or K15-5 (1600V)	Search in LCSC
C9	Capacitor	1000 pF	1	CL-11 (100V) or K10-17 (50V)	Search in LCSC
C10	Capacitor	330 µF	1	Electrolytic 400 V	Search in LCSC
C11, C13	Capacitor	0.47 µF	2	CL-21 (400V) with

Or create a winding, you can assemble a switching type power supply with your own hands, which requires a transformer with only a few turns.

In this case, a small number of parts are required, and the work can be completed in 1 hour. In this case, the IR2151 chip is used as the basis for the power supply.

For work you will need the following materials and parts:

1. PTC thermistor any type.
2. Pair of capacitors, which are selected with the calculation of 1 μF. at 1 W. When creating the design, we select capacitors so that they draw 220 W.
3. Diode assembly"vertical" type.
4. Drivers type IR2152, IR2153, IR2153D.
5. Field effect transistors type IRF740, IRF840. You can choose others if they have a good resistance indicator.
6. Transformer can be taken from old computer system units.
7. Diodes, installed at the outlet, it is recommended to take from the HER family.

In addition, you will need the following tools:

1. Soldering iron and consumables.
2. Screwdriver and pliers.
3. Tweezers.

Also, do not forget about the need for good lighting at the work site.

Step by step instructions

circuit diagram
block diagram

Assembly is carried out according to the drawn circuit diagram. The microcircuit was selected according to the characteristics of the circuit.

Assembly is carried out as follows:

1. At the entrance install a PTC thermistor and diode bridges.
2. Then, a pair of capacitors is installed.
3. Drivers necessary to regulate the operation of the gates of field-effect transistors. If drivers have a D index at the end of the marking, there is no need to install FR107.
4. Field effect transistors installed without shorting the flanges. When attaching to the radiator, use special insulating gaskets and washers.
5. Transformers installed with shorted leads.
6. The output is diodes.

All elements are installed in the designated places on the board and soldered on the reverse side.

Examination

In order to correctly assemble the power supply, you need to be careful about installing the polar elements, and you should also be careful when working with mains voltage. After disconnecting the unit from the power source, there should be no dangerous voltage remaining in the circuit. If assembled correctly, no further adjustment is required.

You can check the correct operation of the power supply as follows:

1. We include in the circuit, at the output of the light bulb, for example, 12 volts. At the first short-term start, the light should be on. In addition, you should pay attention to the fact that all elements should not heat up. If something gets hot, it means the circuit is assembled incorrectly.
2. On the second start We measure the current value using a tester. Let the unit operate for a sufficient amount of time to ensure that there are no heating elements.

In addition, it would be useful to check all elements using a tester for the presence of high current after turning off the power.

1. As previously noted, the operation of a switching power supply is based on feedback. The considered circuit does not require a special organization of feedback and various power filters.
2. Particular attention should be paid to the selection of field-effect transistors. In this case, IR FETs are recommended because they are renowned for their thermal resolution. According to the manufacturer, they can operate stably up to 150 degrees Celsius. However, in this circuit they do not heat up very much, which can be called a very important feature.
3. If the transistors heat up constantly, active cooling should be installed. As a rule, it is represented by a fan.

Advantages and disadvantages

The pulse converter has the following advantages:

1. High rate stabilization coefficient allows you to provide power conditions that will not harm sensitive electronics.
2. Designs considered have a high efficiency rate. Modern versions have this figure at 98%. This is due to the fact that losses are reduced to a minimum, as evidenced by the low heating of the block.
3. Large input voltage range- one of the qualities due to which such a design has spread. At the same time, the efficiency does not depend on the input current indicators. It is the immunity to the voltage indicator that allows you to extend the service life of electronics, since jumps in the voltage indicator are a common occurrence in the domestic power supply network.
4. Input frequency affects the operation of only the input elements of the structure.
5. Small dimensions and weight, are also responsible for their popularity due to the proliferation of portable and portable equipment. After all, when using a linear block, the weight and dimensions increase several times.
6. Organization of remote control.
7. Lower cost.

There are also disadvantages:

1. Availability pulse interference.
2. Necessity inclusion in the circuit of power factor compensators.
3. Complexity self-regulation.
4. Less reliability due to the complexity of the chain.
5. Dire consequences when one or more circuit elements fail.

When creating such a design yourself, you should take into account that mistakes made can lead to failure of the electrical consumer. Therefore, it is necessary to provide protection in the system.

Design and operating features

When considering the operating features of the pulse unit, the following can be noted:

1. At first The input voltage is rectified.
2. Rectified voltage depending on the purpose and features of the entire structure, it is redirected in the form of a high-frequency rectangular pulse and fed to an installed transformer or filter operating at low frequencies.
3. Transformers are small in size and weight when using a pulse unit due to the fact that increasing the frequency makes it possible to increase the efficiency of their operation, as well as reduce the thickness of the core. In addition, ferromagnetic material can be used in the manufacture of the core. At low frequency, only electrical steel can be used.
4. Voltage stabilization occurs through negative feedback. Thanks to the use of this method, the voltage supplied to the consumer remains unchanged, despite fluctuations in the incoming voltage and the generated load.

Feedback can be organized as follows:

1. With galvanic isolation, an optocoupler or transformer winding output is used.
2. If you don't need to create a junction, a resistor voltage divider is used.

Using similar methods, the output voltage is maintained with the required parameters.

Standard switching power supplies, which can be used, for example, to regulate the output voltage during power supply , consists of the following elements:

1. Input part, high voltage. It is usually represented by a pulse generator. Pulse width is the main indicator that affects the output current: the wider the indicator, the greater the voltage, and vice versa. The pulse transformer stands at the section between the input and output parts and separates the pulse.
2. There is a PTC thermistor at the output part. It is made of semiconductor and has a positive temperature coefficient. This feature means that when the temperature of the element increases above a certain value, the resistance indicator increases significantly. Used as a key security mechanism.
3. Low voltage part. The pulse is removed from the low-voltage winding, rectification occurs using a diode, and the capacitor acts as a filter element. The diode assembly can rectify current up to 10A. It should be taken into account that capacitors can be designed for different loads. The capacitor removes the remaining pulse peaks.
4. Drivers carry out quenching of the resistance that arises in the power circuit. During operation, drivers alternately open the gates of installed transistors. Work occurs at a certain frequency
5. Field effect transistors selected taking into account resistance indicators and maximum voltage when open. At a minimum value, the resistance significantly increases efficiency and reduces heating during operation.
6. Transformer standard for downgrade.

Taking into account the chosen circuit, you can begin to create a power supply of the type in question.

Hello everyone!

Background:

On the site there is a diagram of audio frequency power amplifiers (ULF) 125, 250, 500, 1000 Watt, I chose the 500 Watt option, because in addition to radio electronics, I am also a little interested in music and therefore I wanted something of better quality from ULF. The circuit on the TDA 7293 did not suit me, so I decided on the option of 500 watt field-effect transistors. From the beginning I almost assembled one ULF channel, but work stopped for various reasons (time, money and unavailability of some components). As a result, I bought the missing components and completed one channel. Also, after a certain time, I assembled the second channel, set it all up and tested it on a power supply from another amplifier, everything worked at the highest level and I really liked the quality, I didn’t even expect that it would be like this. Special, huge thanks to the radio amateurs Boris, AndReas, Nissan who throughout the entire time I assembled it, helped in setting it up and in other nuances. Next, it became a matter of the power supply. Of course, I would like to make a power supply using a regular transformer, but again everything stops on the availability of materials for the transformer and their cost. Therefore, I decided to stick with the UPS.

Well, now about the UPS itself:

I used IRFP 460 transistors, since I did not find those indicated on the diagram. I had to install the transistors the other way around, turning them 180 degrees, drill more holes for the legs and solder them together with wires (you can see it in the photo). When I made the printed circuit board, I only realized later that I couldn’t find the transistors I needed as in the diagram, so I installed the ones I had (IRFP 460). Transistors and output rectifier diodes must be installed on a heat sink through heat-insulating gaskets, and the radiators must also be cooled with a cooler, otherwise the transistors and rectifier diodes may overheat, but the heating of the transistors, of course, also depends on the type of transistors used. The lower the internal resistance of the field switch, the less it will heat up.

Also, I have not yet installed a 275 Volt Varistor at the input, since it is not in the city and neither is mine, and it is expensive to order one part via the Internet. I will have separate electrolytes at the output, because they are not available for the required voltage and the standard size is not suitable. I decided to put 4 electrolytes of 10,000 microfarads * 50 volts, 2 in series in the arm, in total in each arm it will be 5000 microfarads * 100 volts, which will be completely enough for the power supply, but it is better to put 10,000 microfarads * 100 volts in the shoulder.

The diagram shows a resistor R5 47 kOhm 2 W for powering the microcircuit, it should be replaced with 30 kOhm 5 W (preferably 10 W) so that under a heavy load, the IR2153 microcircuit has enough current, otherwise it may go into protection against a lack of current or will pulsate tension which will affect the quality. In the author’s circuit it is 47 kOhm, which is a lot for such a power supply. By the way, resistor R5 will heat up very much, don’t worry, the type of circuits with IR2151, IR2153, IR2155 power supply is accompanied by strong heating of R5.

In my case, I used an ETD 49 ferrite core and it fit very hard on the board. At a frequency of 56 KHz, according to calculations, it can deliver up to 1400 watts at this frequency, which in my case has a reserve. You can use a toroidal or other shaped core, the main thing is that it is suitable in terms of overall power, permeability and, of course, there is enough space to place it on the board.

Winding data for ETD 49: 1 = 20 turns with 0.63 V wire 5 wires (winding 220 volts). 2 = main power bipolar 2*11 turns of 0.63 V wire 4 wires (winding 2*75-80) volts. 3 = 2.5 turns of wire 0.63 in 1 wire (winding 12 volts, for soft start). 4 = 2 turns of wire 0.63 in 1 wire (additional winding for powering preliminary circuits (timbre block, etc.). The transformer frame needs a vertical design, I have a horizontal one, so I had to fence it. It can be wound in a frameless design. On other types you will have to calculate the core yourself, you can use the program that I will leave at the end of the article. In my case, I used a bipolar voltage of 2 * 75-80 volts for an amplifier of 500 watts, why less, because the amplifier load will not be 8 Ohms but 4 Ohms.

Setup and first launch:

When starting the UPS for the first time, be sure to install a 60-100 watt light bulb in the gap between the network cable and the UPS. When you turn it on, if the light does not light, then it’s good. At the first start-up, short-circuit protection may turn on and the HL1 LED will light up, since the electrolytes have a large capacity and at the moment of switching on take a huge current, if this happens, then you need to twist the multi-turn resistor clockwise until it stops, and then wait until the LED goes out off state and try to turn it on again to make sure the UPS is working, and then adjust the protection. If everything is soldered correctly and the correct part ratings are used, the UPS will start. Next, when you have made sure that the UPS turns on and there is all voltage at the output, you need to set the protection threshold. When setting up protection, be sure to load the UPS between the two arms of the main output winding (which is used to power the ULF) with a 100-watt light bulb. When the HL1 LED lights up when turning on the UPS under load (100-watt light bulb), you need to turn the variable multi-turn resistor R9 2.2 kOhm a little counterclockwise until the power-on protection is triggered. When the LED lights up when turned on, you need to turn it off and wait until it goes out and gradually turn it clockwise in the off state and turn it on again until the protection stops working,
You just need to turn it little by little, for example 1 turn and not 5-10 turns at once, i.e. turned it off, tweaked it and turned it on, the protection worked - again the same procedure several times until you achieve the desired result. When you set the required threshold, then, in principle, the power supply is ready for use and you can remove the mains voltage light and try to load the power supply with an active load, for example, 500 watts. Of course, you can play around with the protection as you like, but I don’t recommend doing tests with Short circuit, since this can lead to a malfunction, even though there is protection, some capacity will not have time to discharge, the relay will not respond instantly or will stick and there may be trouble. Although I accidentally and not accidentally made a number of short circuits, the protection works. But nothing is eternal.

DIY PULSE POWER SUPPLY FOR IR2153

Functionally, the IR2153 microcircuits differ only in the Voltage Booster diode installed in the planar housing:

Functional diagram of IR2153

Functional diagram of IR2153D

First, let's look at how the microcircuit itself works, and only then we will decide which power supply to assemble from it. First, let's look at how the generator itself works. The figure below shows a fragment of a resistive divider, three op-amps and an RS trigger:

At the initial moment of time, when the supply voltage has just been applied, capacitor C1 is not charged at all inverting inputs of the op-amp there is a zero, and at the non-inverting inputs there is a positive voltage generated by a resistive divider. As a result, it turns out that the voltage at the inverting inputs is less than at the non-inverting inputs and all three op-amps at their outputs generate a voltage close to the supply voltage, i.e. log unit.
Since the input R (zero setting) on ​​the trigger is inverting, then for it this will be a state in which it does not affect the state of the trigger, but at the input S there will be a log of one, which sets the output of the trigger to also a log of one and a capacitor Ct through resistor R1 will start charging. In the picture the voltage across Ct is shown by the blue line,red - voltage at output DA1, green - at output DA2, A pink - at the output of the RS trigger:

As soon as the voltage on Ct exceeds 5 V, a log zero is formed at the DA2 output, and when, continuing to charge Ct, the voltage reaches a value slightly more than 10 volts, a log zero will appear at the DA1 output, which in turn will serve to set the RS trigger to the log zero state. From this moment, Ct will begin to discharge, also through resistor R1, and as soon as the voltage on it becomes slightly less than the set value by dividing the value of 10 V, a log unit will again appear at the output of DA1. When the voltage on the capacitor Ct becomes less than 5 V, a log one will appear at the output of DA2 and turn the RS trigger into the one state and Ct will begin to charge again. Of course, at the inverse output of the RS trigger, the voltage will have opposite logical values.
Thus, at the outputs of the RS trigger, levels of log one and zero are formed that are opposite in phase, but equal in duration:

Since the duration of the control pulses IR2153 depends on the charge-discharge rate of the capacitor Ct, it is necessary to carefully pay attention to flushing the board from flux - there should be no leaks from the terminals of the capacitor or from the printed conductors of the board, since this is fraught with magnetization of the core of the power transformer and failure power transistors.
There are also two more modules in the chip - UV DETECT And LOGIK. The first of them is responsible for starting and stopping the generating process, depending on the supply voltage, and the second generates pulses DEAD TIME, which are necessary to eliminate the through current of the power stage.
Next, the logical levels are separated - one becomes the control upper arm of the half-bridge, and the second the lower one. The difference is that the upper arm is controlled by two field-effect transistors, which, in turn, control the final stage, which is “detached” from the ground and “detached” from the supply voltage. If we consider a simplified circuit diagram for connecting the IR2153, it turns out something like this:

Pins 8, 7 and 6 of the IR2153 microcircuit are outputs VB, HO and VS, respectively, i.e. power supply for the upper side control, the output of the final stage of the upper side control and the negative wire of the upper side control module. Attention should be paid to the fact that at the moment of switching on, the control voltage is present at the Q RS trigger, therefore the low-side power transistor is open. Capacitor C3 is charged through diode VD1, since its lower terminal is connected to the common wire through transistor VT2.
As soon as the RS trigger of the microcircuit changes its state, VT2 closes, and the control voltage at pin 7 of IR2153 opens transistor VT1. At this moment, the voltage at pin 6 of the microcircuit begins to increase, and to keep VT1 open, the voltage at its gate must be greater than at the source. Since the resistance of an open transistor is equal to tenths of an ohm, the voltage at its drain is not much greater than at the source. It turns out that holding the transistor in the open state requires a voltage of at least 5 volts greater than the supply voltage, and indeed it is - capacitor C3 is charged to 15 volts and it is this that allows you to keep VT1 in the open state, since the energy stored in it during this the moment of time is the supply voltage for the upper arm of the window stage of the microcircuit. Diode VD1 at this point in time does not allow C3 to discharge to the power bus of the microcircuit itself.
As soon as the control pulse at pin 7 ends, transistor VT1 closes and then VT2 opens, which again charges capacitor C3 to a voltage of 15 V.

Quite often, amateurs install an electrolytic capacitor with a capacity of 10 to 100 μF in parallel with capacitor C3, without even delving into the need for this capacitor. The fact is that the microcircuit is capable of operating at frequencies from 10 Hz to 300 kHz and the need for this electrolyte is relevant only up to frequencies of 10 kHz, and then only on condition that the electrolytic capacitor is of the WL or WZ series - technologically they have a small ers and are better known as computer capacitors with inscriptions in gold or silver paint:

For popular conversion frequencies used in the creation of switching power supplies, frequencies are taken above 40 kHz, and sometimes raised to 60-80 kHz, so the relevance of using an electrolyte simply disappears - even a capacitance of 0.22 μF is already enough to open and hold the SPW47N60C3 transistor open , which has a gate capacitance of 6800 pF. To ease the conscience, a 1 µF capacitor is installed, and allowing for the fact that IR2153 cannot switch such powerful transistors directly, the accumulated energy of capacitor C3 is enough to control transistors with a gate capacitance of up to 2000 pF, i.e. all transistors with a maximum current of about 10 A (the list of transistors is below in the table). If you still have doubts, then instead of the recommended 1 µF, use a 4.7 µF ceramic capacitor, but this is pointless:

It would be unfair not to note that the IR2153 microcircuit has analogues, i.e. microcircuits with a similar functional purpose. These are IR2151 and IR2155. For clarity, let’s put the main parameters in a table, and then we’ll figure out which of them is best to prepare:

CHIP	Maximum Driver Voltage	Start supply voltage	Stop supply voltage	Maximum current for charging the gates of power transistors / rise time	Maximum power transistor gate discharge current/fall time	Internal Zener diode voltage
				100 mA / 80...120 nS	210 mA / 40...70 nS
				NOT SPECIFIED / 80...150 nS	NOT SPECIFIED / 45...100 nS
				210 mA / 80...120 nS	420 mA / 40...70 nS

As can be seen from the table, the differences between the microcircuits are not very large - all three have the same shunt zener diode power supply, the start and stop supply voltages are almost the same for all three. The difference lies only in the maximum current of the final stage, which determines which power transistors and at what frequencies the microcircuits can control. Oddly enough, the most hyped IR2153 turned out to be neither fish nor fowl - it does not have a standardized maximum current of the last driver stage, and the rise-fall time is somewhat prolonged. They also differ in cost - IR2153 is the cheapest, but IR2155 is the most expensive.
The generator frequency is the conversion frequency ( no need to divide by 2) for IR2151 and IR2155 is determined by the formulas given below, and the frequency of IR2153 can be determined from the graph:

In order to find out which transistors can be controlled by the IR2151, IR2153 and IR2155 microcircuits, you should know the parameters of these transistors. The greatest interest when connecting a microcircuit and power transistors is the gate energy Qg, since it is this energy that will influence the instantaneous values ​​of the maximum current of the microcircuit drivers, which means a table with transistor parameters will be required. Here SPECIAL Attention should be paid to the manufacturer, since this parameter differs from different manufacturers. This is most clearly seen in the example of the IRFP450 transistor.
I understand perfectly well that for a one-time production of a power supply, ten to twenty transistors are still too much, nevertheless, I posted a link for each type of transistor - I usually buy there. So click, see the prices, compare with retail and the likelihood of buying lefty. Of course, I’m not saying that on Ali there are only honest sellers and all the goods are of the highest quality - there are a lot of scammers everywhere. However, if you order transistors that are produced directly in China, it is much more difficult to run into crap. And it is for this reason that I prefer STP and STW transistors, and I don’t even hesitate to buy them from disassembly, i.e. Used.

POPULAR TRANSISTORS FOR PULSE POWER SUPPLY
NAME	VOLTAGE			POWER	CAPACITY SHUTTER	Qg (PRODUCER)
NETWORK (220 V)
						17...23nC ( ST)
						38...50nC ( ST)
						35...40nC ( ST)
						39...50nC ( ST)
						46nC ( ST)
						50...70nC ( ST)
						75nC ( ST)
						84nC ( ST)
						65nC ( ST)
						46nC ( ST)
						50...70nC ( ST)
						75nC ( ST)
						65nC ( ST)
STP20NM60FP						54nC ( ST)
						150nC(IR) 75nC ( ST)
						150...200nC (IN)
						252...320nC (IN)
						87...117nC ( ST)

I g = Q g / t on = 63 x 10 -9 / 120 x 10 –9 = 0.525 (A) (1)

When the amplitude of the control voltage pulses at the gate is Ug = 15 V, the sum of the output resistance of the driver and the resistance of the limiting resistor should not exceed:

Rmax = U g / I g = 15 / 0.525 = 29 (Ohm) (2)

Let's calculate the output impedance of the driver stage for the IR2155 chip:

R on = U cc / I max = 15V / 210mA = 71.43 ohms
R off = U cc / I max = 15V / 420mA = 33.71 ohms

Taking into account the calculated value according to formula (2) Rmax = 29 Ohm, we come to the conclusion that with the IR2155 driver it is impossible to achieve the specified speed of the IRF840 transistor. If a resistor Rg = 22 Ohm is installed in the gate circuit, the turn-on time of the transistor will be determined as follows:

RE on = R on + R gate, where RE - total resistance, R R gate - resistance installed in the gate circuit of the power transistor = 71.43 + 22 = 93.43 ohms;
I on = U g / RE on, where I on is the opening current, U g - gate control voltage value = 15 / 93.43 = 160mA;
t on = Q g / I on = 63 x 10-9 / 0.16 = 392nS
The shutdown time can be calculated using the same formulas:
RE off = R out + R gate, where RE - total resistance, R out - driver output impedance, R gate - resistance installed in the gate circuit of the power transistor = 36.71 + 22 = 57.71 ohms;
I off = U g / RE off, where I off - opening current, U g - gate control voltage value = 15 / 58 = 259mA;
t off = Q g / I off = 63 x 10-9 / 0.26 = 242nS
To the resulting values ​​it is necessary to add the time of the transistor’s own opening and closing, resulting in the real time t on will be 392 + 40 = 432nS, and t off 242 + 80 = 322nS.
Now all that remains is to make sure that one power transistor has time to close completely before the second one begins to open. To do this, add t on and off getting 432 + 322 = 754 nS, i.e. 0.754 µS. What is this for? The fact is that any of the microcircuits, be it IR2151, or IR2153, or IR2155, has a fixed value DEAD TIME, which is 1.2 µS and does not depend on the frequency of the master oscillator. The datasheet mentions that Deadtime (typ.) 1.2 µs, but it also contains a very confusing drawing from which the conclusion suggests itself that DEAD TIME is 10% of the control pulse duration:

To dispel doubts, the microcircuit was turned on and a two-channel oscilloscope was connected to it:

The power supply was 15 V, and the frequency was 96 kHz. As can be seen from the photograph, with a scan of 1 µS, the duration of the pause is quite a bit more than one division, which exactly corresponds to approximately 1.2 µS. Next we reduce the frequency and see the following:

As can be seen from the photo, at a frequency of 47 kHz, the pause time has practically not changed, therefore the sign stating that Deadtime (typ.) 1.2 µs is true.
Since the microcircuits were already working, it was impossible to resist one more experiment - lowering the supply voltage to make sure that the generator frequency would increase. The result is the following picture:

However, expectations were not met - instead of increasing the frequency, it decreased, by less than 2%, which can be generally ignored and noted that the IR2153 microcircuit keeps the frequency quite stable - the supply voltage has changed by more than 30%. It should also be noted that the pause time has increased slightly. This fact is somewhat pleasing - as the control voltage decreases, the opening and closing time of the power transistors increases slightly and increasing the pause in this case will be very useful.
It was also found that UV DETECT copes with its function perfectly - with a further decrease in the supply voltage, the generator stopped, and with an increase, the microcircuit started again.
Now let’s return to our mathematics, based on the results of which we found that with 22 Ohm resistors installed in the gates, the closing and opening time is equal to 0.754 µS for the IRF840 transistor, which is less than the 1.2 µS pause given by the microcircuit itself.
Thus, with the IR2155 microcircuit through 22 Ohm resistors it will be quite normal to control the IRF840, but the IR2151 will most likely have a long life, since to close and open the transistors we needed a current of 259 mA and 160 mA, respectively, and its maximum values ​​are 210 mA and 100 ma. Of course, you can increase the resistance installed in the gates of power transistors, but in this case there is a risk of going beyond the limits DEAD TIME. In order not to engage in fortune telling on coffee grounds, a table was compiled in EXCEL, which you can take. It is assumed that the supply voltage of the microcircuit is 15 V.
To reduce switching noise and slightly reduce the closing time of power transistors in switching power supplies, either the power transistor is shunted with a resistor and capacitor connected in series, or the power transformer itself is shunted with the same chain. This node is called a snubber. The snubber circuit resistor is chosen with a value 5–10 times greater than the drain-source resistance of the field-effect transistor in the open state. The capacitance of the circuit capacitor is determined from the expression:
C = tdt/30 x R
where tdt is the pause time for switching the upper and lower transistors. Based on the fact that the duration of the transient process, equal to 3RC, should be 10 times less than the duration of the dead time value tdt.
Damping delays the opening and closing moments of the field-effect transistor relative to differences in the control voltage across its gate and reduces the rate of change in voltage between the drain and the gate. As a result, the peak values ​​of the flowing current pulses are smaller and their duration is longer. Almost without changing the turn-on time, the damping circuit noticeably reduces the turn-off time of the field-effect transistor and limits the spectrum of generated radio interference.

Now that we've sorted out the theory a little, we can move on to practical schemes.
The simplest switching power supply circuit based on IR2153 is an electronic transformer with a minimum of functions:

The circuit does not have any additional functions, and the secondary bipolar power supply is formed by two rectifiers with a midpoint and a pair of dual Schottky diodes. The capacitance of capacitor C3 is determined at the rate of 1 μF of capacitance per 1 W of load. Capacitors C7 and C8 are of equal capacity and range from 1 µF to 2.2 µF. The power depends on the core used and the maximum current of the power transistors and theoretically can reach 1500 W. However, this is only THEORETICALLY , based on the fact that 155 VAC is applied to the transformer, and the maximum current of the STP10NK60Z reaches 10A. In practice, all datasheets indicate a decrease in the maximum current depending on the temperature of the transistor crystal, and for the STP10NK60Z transistor the maximum current is 10 A at a crystal temperature of 25 degrees Celsius. At a crystal temperature of 100 degrees Celsius, the maximum current is already 5.7 A and we are talking specifically about the temperature of the crystal, and not the heat sink flange, and even more so about the temperature of the radiator.
Therefore, the maximum power should be selected based on the maximum current of the transistor divided by 3 if it is a power supply for a power amplifier and divided by 4 if it is a power supply for a constant load, such as incandescent lamps.
Considering the above, we find that for a power amplifier you can get a switching power supply with a power of 10 / 3 = 3.3A, 3.3A x 155V = 511W. For a constant load we get a power supply 10/4 = 2.5 A, 2.5 A x 155V = 387W. In both cases, 100% efficiency is used, which does not happen in nature. In addition, if we assume that 1 µF of the primary power supply capacity per 1 W of load power, then we will need a capacitor, or capacitors with a capacity of 1500 µF, and such a capacitance must be charged through soft start systems.
A switching power supply with overload protection and soft start via secondary power supply is presented in the following diagram:

First of all, this power supply has overload protection made on the current transformer. You can read details about calculating a current transformer. However, in the vast majority of cases, a ferrite ring with a diameter of 12...16 mm, on which about 60...80 turns are wound in two wires, is quite sufficient. Diameter 0.1...0.15 mm. Then the beginning of one winding is connected to the ends of the second. This is the secondary winding. The primary winding contains one or two, sometimes one and a half turns are more convenient.
Also in the circuit, the values ​​of resistor R4 and R6 are reduced in order to expand the range of the primary supply voltage (180...240V). In order not to overload the zener diode installed in the microcircuit, the circuit has a separate zener diode with a power of 1.3 W at 15 V.
In addition, a soft start for secondary power was introduced into the power supply, which made it possible to increase the capacitance of the secondary power filters to 1000 µF at an output voltage of ±80 V. Without this system, the power supply entered protection at the moment of switching on. The principle of operation of the protection is based on the operation of IR2153 at an increased frequency at the moment of switching on. This causes losses in the transformer and it is not able to deliver maximum power to the load. As soon as generation begins through the divider R8-R9, the voltage supplied to the transformer reaches the detector VD5 and VD7 and charging of the capacitor C7 begins. As soon as the voltage becomes sufficient to open VT1, C3 is connected to the frequency-setting chain of the microcircuit and the microcircuit reaches the operating frequency.
Additional inductances for the primary and secondary voltages have also been introduced. Inductance on the primary power supply reduces interference created by the power supply and going into the 220V network, and on the secondary power supply it reduces RF ripple on the load.
In this version there are two additional secondary supplies. The first is intended to power a twelve-volt computer cooler, and the second is to power the preliminary stages of a power amplifier.
Another sub-option of the circuit is a switching power supply with a unipolar output voltage:

Of course, the secondary winding is designed for the voltage that is needed. The power supply can be soldered on the same board without mounting elements that are not on the diagram.

The next version of the switching power supply is capable of delivering about 1500 W to the load and contains soft start systems for both primary and secondary power, has overload protection and voltage for the forced cooling cooler. The problem of controlling powerful power transistors is solved by using emitter followers on transistors VT1 and VT2, which discharge the gate capacitance of powerful transistors through themselves:

Such forcing of the closing of power transistors allows the use of quite powerful specimens, such as IRFPS37N50A, SPW35N60C3, not to mention IRFP360 and IRFP460.
At the moment of switching on, the voltage is supplied to the primary power diode bridge through resistor R1, since the contacts of relay K1 are open. Next, the voltage is supplied through R5 to the microcircuit and through R11 and R12 to the output of the relay winding. However, the voltage increases gradually - C10 has a fairly large capacity. From the second winding of the relay, voltage is supplied to the zener diode and thyristor VS2. As soon as the voltage reaches 13 V, it will be enough to pass through the 12-volt zener diode to open VS2. Here it should be recalled that IR2155 starts with a supply voltage of approximately 9 V, therefore, at the time of opening, VS2 will already generate control pulses through IR2155, only they will enter the primary winding through resistor R17 and capacitor C14, since the second group of contacts of relay K1 is also open . This will significantly limit the charging current of the secondary power filter capacitors. As soon as the thyristor VS2 opens, voltage will be applied to the relay winding and both contact groups will close. The first will bypass the current-limiting resistor R1, and the second - R17 and C14.
The power transformer has a service winding and a rectifier on diodes VD10 and VD11 from which the relay will be powered, as well as additional power supply to the microcircuit. R14 serves to limit the forced cooling fan current.
Thyristors used VS1 and VS2 - MCR100-8 or similar in TO-92 housing
Well, at the end of this page, another circuit is still on the same IR2155, but this time it will act as a voltage stabilizer:

As in the previous version, the power transistors are closed by bipolars VT4 and VT5. The circuit is equipped with a soft start of the secondary voltage on VT1. The start is made from the vehicle’s on-board power supply and then the power is supplied by a stabilized voltage of 15 V, vortexed by diodes VD8, VD9, resistor R10 and zener diode VD6.
There is another rather interesting element in this circuit - tC. This is heatsink overheat protection that can be used with almost any converter. It was not possible to find an unambiguous name; in common parlance it is a self-restoring thermal fuse; in price lists it is usually designated KSD301. It is used in many household electrical appliances as a protective or temperature-regulating element, since they are produced with different response temperatures. This fuse looks like this:

As soon as the radiator temperature reaches the fuse cut-off limit, the control voltage from the REM point will be removed and the converter will turn off. After the temperature drops by 5-10 degrees, the fuse will be restored and supply control voltage and the converter will start again. The same thermal fuse, or thermal relay, can also be used in network power supplies by monitoring the temperature of the radiator and turning off the power, preferably low-voltage, going to the microcircuit - the thermal relay will work longer this way. You can buy KSD301.
VD4, VD5 - fast diodes from the SF16, HER106, etc. series.
Overload protection can be introduced into the circuit, but during its development the main emphasis was on miniaturization - even the soft start unit was a big question.
The manufacture of winding parts and printed circuit boards are described on the following pages of the article.

Well, at the end of the day there are several circuits of switching power supplies found on the Internet.
Scheme No. 6 taken from the SOLDERING IRON website:

In the next power supply on the self-clocked driver IR2153, the capacitance of the boost capacitor is reduced to a minimum of 0.22 μF (C10). The microcircuit is powered from an artificial midpoint of the power transformer, which is not important. There is no overload protection; the shape of the voltage supplied to the power transformer is slightly corrected by the inductance L1:

While selecting diagrams for this article, I came across this one. The idea is to use two IR2153 in a bridge converter. The author's idea is quite clear - the output of the RS trigger is fed to the input Ct and, according to logic, control pulses of opposite phase should be generated at the outputs of the slave microcircuit.
The idea intrigued me and an investigative experiment was carried out on the topic of testing its functionality. It was not possible to obtain stable control pulses at the outputs of IC2 - either the upper driver or the lower one was working. In addition, the pause phase changed DEAD TIME, on one microcircuit relative to another, which will significantly reduce the efficiency and the idea was forced to be abandoned.

The distinctive feature of the next power supply on the IR2153 is that if it works, then this work is akin to a powder keg. First of all, the additional winding on the power transformer to power the IR2153 itself caught my eye. However, there is no current-limiting resistor after diodes D3 and D6, which means that the fifteen-volt zener diode located inside the microcircuit will be VERY heavily loaded. One can only guess what will happen if it overheats and undergoes thermal breakdown.
The overload protection on VT3 bypasses the time setting capacitor C13, which is quite acceptable.

The last acceptable version of the power source circuit on the IR2153 does not represent anything unique. True, for some reason the author too reduced the resistance of the resistors in the gates of the power transistors and installed zener diodes D2 and D3, the purpose of which is not very clear. In addition, the capacitance C11 is too small, although perhaps we are talking about a resonant converter.

There is another option for a switching power supply using IR2155 and specifically for controlling a bridge converter. But there the microcircuit controls power transistors through an additional driver and matching transformer, and we are talking about induction melting of metals, so this option deserves a separate page, and everyone who understands at least half of what you read should go to the page with PRINTED BOARDS.

VIDEO INSTRUCTIONS FOR SELF ASSEMBLY
SWITCH POWER SUPPLY BASED ON IR2153 OR IR2155

A few words about the manufacture of pulse transformers:

How to determine the number of turns without knowing the grade of ferrite: